Method and system for moving target elimination and indication using smoothing filters

ABSTRACT

In-phase sum and difference signals and quadrature sum and difference signals from a monopulse radar system are processed to form the sum of the in-phase signals, the sum of the quadrature signals, the difference of the in-phase signals, and the difference of the quadrature signals. The processed sum signals and the processed difference signals are then combined to form complex signals one of which is advanced and the other retarded by pedetermined time. The complex signals are then divided into sequences of frequencies of identical banks of narrow band filters. The differences between the outputs of corresponding filters from each bank are smoothed by low pass filters, averaged and then divided by a constant to form a beam pointing error value.

BACKGROUND OF THE INVENTION

This invention relates to radar for air launched missile systems, andmore particularly to a null command generator moving target eliminatorand indicator.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present invention is an improvement over my copending applicationSer. No. 233,835 filed herewith and entitled, "Method and System forMoving Target Elimination and Indication".

The improved invention is a novel system for processing radar reflectionfrom a monopulse antenna system and converting these signals into beampointing error signals which can be used to alter the position of theantenna and thereby track a target. The process also acts as a movingtarget indicator.

The invention performs two primary functions, in that it reduces theeffects of moving targets on a null command generator, and it indicatesthe presence of moving targets. It is in effect a novel null commandgenerator that incorporates some of the principles of what is referredto as an advanced null command generator.

The improved processor by a smoothing operation of phase differencesreduces its sensitivity to noise.

SUMMARY OF THE INVENTION

The invention presented here processes input signals derived from amonopulse radar system, these signals including in-phase and quadraturesum signals and in-phase and quadrature difference signals.Preprocessing converts these signals into complex signals A and B whereA=A_(in) +jA_(qd) and B=B_(in) +jB_(qd) and A and B are then delayed andadvanced respectively. Each is fed through identical banks of narrowband filters and the phase difference of corresponding filters of eachbank are smoothed in low pass filters and averaged. The purpose of thesmoothing is to reduce the errors in the individual phase estimatesbefore selective averaging takes place.

It is an object of the invention to provide a novel and improved radarprocessor.

It is another object to provide a null command generator system forprocessing radar reflections from a monopulse radar for an air launchedmissile.

It is another object to provide a monopulse radar system that reducesthe effect of moving targets on a null command generator and alsoindicates their presence.

These and other advantages, features and objects of the invention willbecome more apparent from the following description taken in connectionwith the illustrative embodiment in the accompanying drawings.

DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing the conventional input to a nullcommand generator;

FIG. 2 is a block diagram showing the preprocessor for the null commandgenerator; and

FIG. 3 is a block diagram showing the null command generator with theadditional smoothing filters.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

The input to the improved null command generator is the same as thatused in the past and is shown in FIG. 1. The radar signals are receivedin monopulse antennas 11 and 12 and are fed to magic tee circuit 15which forms a sum and difference that are fed respectively to mixers 17and 18 which are also fed by local oscillator 21 which serves assynthetic target signals. Mixers 17 and 18 are then fed to intermediatefrequency amplifiers 23 and 24 respectively which are then fed toin-phase and quadrature detectors 27 and 28. Detector 27 producesin-phase and quadrature sum signals while detector 28 produces in-phaseand quadrature difference signals. The sum and difference of thein-phase signals are formed respectively in adder 31 and subtractor 32while the sum and difference of the quadrature signals are formedrespectively in adder 35 and subtractor 36. The sums of the outputs ofsubtracting circuits 33 and 36 and adder circuits 33 and 35 are thenrespectively formed in adders 39 and 40 and produce signals A and Bwhich are advanced and retarded in circuits 43 and 44 by τ_(d) which isequal to the difference between the antenna phase centers divided by theproduct of four times the velocity of the aircraft and the sine of thetarget azimuths. The outputs of circuits 43 and 44 produce signals A_(d)and B_(a).

As shown in FIG. 2, the processor now passes the A_(d) and B_(a) signalsthrough identical narrow band filter banks 47 and 48 having n filters.The narrow band filter outputs are processed in various ways. First, thephase difference between the narrow band filter A₁ and the narrow bandfilter B₁ is formed in circuit 51. This is proportional to the antennabeam pointing error; that is, the angle between the monopulse null planeand the synthetic target. The phase angles between the other A_(i) andB_(i) complex quantities are similarly computed in circuits 52 to 53.

The improvement over the processor disclosed in my previously filedapplication mentioned aforesaid is the inclusion here of smoothingfilters 51a, 52a, and 53 between the φ_(i) estimators and the selectiveaveraging. The purpose of this smoothing is to reduce the errors in theindividual phase estimates before selective averaging takes place. Thisreduces the number of φ_(i) estimates that are erroneously rejected oraccepted in the selective averaging. The smoothing filter can beconsidered as a low pass or an averaging over the past history of theparticular φ_(i).

The outputs of the smoothing filters are indicated as φ_(i).

The modified processor of this report disclosed here gives betterperformance than does the processor without the smoothing filters. Thisis due to its reduced sensitivity to noise. As before the beam pointingerror output, ε_(n) and the moving target output indication outputs canbe further smoothed.

The first average of the previously computed phase angles φ_(i) iscomputed in the selective averaging circuit 55. This average is notφ_(a). Before φ_(a) is computed the absolute value of the differencebetween each φ_(i) and this first average is compared with a constant. Asecond φ_(i) average is now computed using only those φ_(i) whoseabsolute difference from the first mean did not exceed the constant.This second average then is denoted φ_(a).

The A_(i) outputs of the A_(d) channel narrow band filter bank are nowadvanced in phase by an amount φ_(a) in phase advance circuits 57. Next,the difference of these signals and their correspondents that are notphase rotated in the B_(i) channels is formed in circuits 59, and thenthe absolute value squared denoted as |R_(i) |² of each of thesedifferences is formed in circuits 61. They are greater when movingtarget signals are present in the corresponding narrow band filters.

In order to compute a reference for each of the above, the absolutevalues squared of the output of each narrow band filter are formed incircuits 63 and 64. Corresponding absolute squared values of the A and Bchannels are added in adder 62 and multiplied by α, a fixed constant, inmultiplier 65 forming a reference value or threshold.

The previously computed |R_(i) |² quantities are now compared with theirreferences in threshold circuit 67 by subtraction thereby performing therelation

    |R.sub.i |.sup.2 -α{|A.sub.i |.sup.2 +|B.sub.i |.sup.2 }.

If the fixed positive quantity is exceeded by any channels, then theφ_(i) of those channels are not used in computing the final phase,φ_(out).

The final phase estimate is accomplished by taking an average of theφ_(i) from those channels whose |R_(i) |² were sufficiently small, andconsequently did not have a threshold crossing as just described. Thisis again carried out in the selective averaging circuit with selectorcontrol 69 controlling which φ_(i) 's are to be added.

To obtain the estimate of the beam pointing error the φ_(out) quantityis divided by the constant (πl.sub.φ /λ) in divisor 71 where l.sub.φ isthe distance between antenna phase center 71. The result is an estimateof the beam pointing error.

Many modifications of the above null command generator processor arepossible. For example, the φ_(i) estimates selected for rejection neednot be determined each time new A_(i) and B_(i) are determined. Pastrejection information can be used and periodically updated. The beampointing estimates can be smoothed, or the φ_(out) estimates can besmoothed before the division process. Also, the threshold circuits andthe related computations can be omitted. In this case φ_(a) is taken asφ_(out). This concept, as presented in this invention, is alsoapplicable to synthetic array and doppler beam sharpening systems toindicate moving targets.

What is claimed is:
 1. In a monopulse radar system, a null commandgenerator fed by the in-phase and quadrature sum and difference signalsderived from target reflections received at monopulse antennascomprising:a. a first subtracting circuit fed by the in-phase sum anddifference signals; b. a second subtracting circuit fed by thequadarature sum and difference signals; c. a first summing circuit fedby the in-phase sum and difference signals; d. a second summing circuitfed by the quadrature sum and difference signals; e. a third summingcircuit fed by the first and second subtracting circuits; f. a fourthsumming circuit fed by the first and second summing circuits; g. meansfor delaying the output of the third summing circuit by a time valueτ_(d), where τ_(d) is a value equal to the distance between the antennaphase center divided by the product of 4 times the product of theaircraft velocity and the target azimuth; h. means for advancing theoutput of the fourth summing circuit by τ_(d) ; i. a first bank ofnarrow band filters fed by the delaying means; j. a second bank ofnarrow band filters identical to that of the first bank fed by theadvancing means; k. means for phase differencing corresponding filtersof the first and second banks of narrow band filters; l. a plurality ofmeans for smoothing the outputs of the phase differencing means; and m.means for averaging the outputs of the smoothing means.
 2. A nullcommand generator according to claim 1 which further comprises:a. a bankof means for advancing the phase of the output for a first bank ofnarrow band filters; b. a bank of subtracting circuits fed bycorresponding outputs of the advancing means bank and the second bank ofnarrow band filters; c. a first bank of means for obtaining the squareof the absolute value of the outputs of the bank of subtractingcircuits; d. a second bank of means for obtaining the square of theabsolute value of the outputs of the first bank of narrow bandfilters;e. a third bank of means for obtaining the square of theabsolute value of the outputs of the second bank of filters; f. meansfor adding the outputs of corresponding absolute value squaring means ofthe second and third banks thereof; g. a bank of means for multiplying aconstant to the output of the bank of adding means forming a thresholdsignal; h. means or comparing the threshold signals with thecorresponding output of the first bank of absolute squaring means forselection control, the output of the comparing means being fed to theaveraging means; and i. means for dividing the output of the averagingmeans by a value equal to the product of πtimes the distance between thephase centers of the antennas divided by the wavelength of the radarsignal.
 3. A null command generator according to claim 2 wherein thesmoothing means comprises low pass filters.
 4. A method of generating anerror signal from inphase and quadrature sum signals and in-phase andquadrature difference signals derived from a monopulse radar systemcomprising:a. subtracting the in-phase signals forming a firstdifference; b. adding the in-phase signals forming a first sum; c.subtracting the quadrature signals forming a second difference; d.adding the quadrature signals forming a second sum; e. adding the firstand second sums forming a first complex signal; f. adding the first andsecond differences forming a second complex signal; g. advancing thefirst complex signal by a predetermined value; h. delaying the secondcomplex signal by said predetermined value; i. narrow band filtering thedelayed complex signal into a first sequence of frequency bands; j.narrow band filtering the advanced complex signal into a second sequenceof frequency bands corresponding to the first sequence; k. phasedifferencing the signals from corresponding frequency bands of the firstand second sequences forming a sequence of phase difference signals; l.smoothing each of the sequence of the phase difference signals; and m.averaging the smoothed phase difference signals.
 5. A method ofgenerating an error signal according to claim 4 which furtherincludes:a. advancing the phase of the first sequence by a value equalto the phase difference average; b. subtracting the second sequence fromthe advanced first sequence forming a sequence of phase rotated signals;c. squaring the absolute value of the sequence of phase rotated signals;d. squaring the absolute value of the first sequence of the frequencyband; e. squaring the absolute value of the second sequence of thefrequency band; f. adding the corresponding absolute value squared ofthe first and second sequences to form a sequence of absolute squaredsums; g. multiplying each of the absolute squared sums by a constant toform a sequence of thresholds; h. subtracting the sequence of thresholdsfrom the sequence of phase rotated signals; and i. averaging thesequence of threshold signal differences that exceed a predeterminedvalue.
 6. A method of generating an error signal according to claim 5wherein the phase difference smoothing comprises low pass filtering.